Predistortion Circuit, Method For Generating A Predistorted Baseband Signal, Control Circuit For A Predistortion Circuit, Method To Determine Parameters For A Predistortion Circuit, And Apparatus And Method For Predistorting A Baseband Signal

ABSTRACT

A predistortion circuit for a wireless transmitter includes a signal input configured to receive a baseband signal. Further, the predistortion circuit includes a predistorter configured to generate a predistorted baseband signal using the baseband signal and a select of one of a first predistorter configuration and a second predistorter configuration.

PRIORITY CLAIM

This application is a divisional of U.S. patent application Ser. No.16/766,416, filed May 22, 2020, titled “Predistortion Circuit, MethodFor Generating A Predistorted Baseband Signal, Control Circuit For APredistortion Circuit, Method To Determine Parameters For APredistortion Circuit, And Apparatus And Method For Predistorting ABaseband Signal,” which is a U.S. National Stage filing of InternationalApplication No. PCT/US2017/068859, filed Dec. 29, 2017, titled“Predistortion Circuit, Method For Generating A Predistorted BasebandSignal, Control Circuit For A Predistortion Circuit, Method To DetermineParameters For A Predistortion Circuit, And Apparatus And Method ForPredistorting A Baseband Signal,” all of which are incorporated hereinby reference in their entirety.

The claims in the instant application are different than those of theparent application and/or other related applications. The Applicanttherefore rescinds any disclaimer of claim scope made in the parentapplication and/or any predecessor application in relation to theinstant application. Any such previous disclaimer and the citedreferences that it was made to avoid, may need to be revisited. Further,any disclaimer made in the instant application should not be read intoor against the parent application and/or other related applications.

FIELD

The present disclosure relates to predistortion circuits for wirelesstransmitters and to control circuits for the predistortion circuits.

BRIEF DESCRIPTION OF THE FIGURES

Some examples of apparatuses and/or methods will be described in thefollowing by way of example only, and with reference to the accompanyingfigures, in which

FIG. 1 illustrates an example of a predistortion circuit for a wirelesstransmitter;

FIG. 2 illustrates an example of a predistortion circuit within aTransmitter;

FIG. 3 illustrates an example for operation characteristics that may beused to select between at least two predistorter characteristics;

FIG. 4 illustrates a flowchart of an example of a method for generatinga predistorted baseband signal;

FIG. 5 illustrates an example of a predistortion circuit havingcomputation nodes operating at different rates;

FIG. 6 illustrates a further example of a predistortion circuit havingcomputation nodes operating at different rates;

FIG. 7 illustrates aliasing deteriorations of a predistorted basebandsignal which may be mitigated by an example of a predistortion circuit;

FIG. 8 illustrates an example of a control circuit;

FIG. 9 illustrates an example of a control circuit within a transmitter;

FIG. 10 illustrates an example of a signal spectrum;

FIG. 11 illustrates an example of a control circuit configured tocompare bandwidth limited portions of a reference signal and of afeedback signal;

FIG. 12 illustrates an overview of a method to compare a referencesignal 1210 and a feedback signal 1220;

FIG. 13 illustrates a graphical representation of the generation of thebandlimited signals according to FIG. 11;

FIG. 14 illustrates an example of a millimeter wave implementation;

FIG. 15 illustrates a system of system of equations to be optimized todetermine predistorter parameters;

FIG. 16 illustrates an example of an apparatus for predistorting abaseband signal;

FIG. 17 illustrates a signal sampled at a first sample rate;

FIG. 18 illustrates a signal sampled at a second sample rate;

FIG. 19 illustrates a predistorted baseband signal;

FIGS. 20 to 22 illustrate further examples of signal spectra;

FIG. 23 illustrates an example of a wireless transceiver;

FIG. 24 illustrates a flowchart of an example of a method forpredistorting a baseband signal; and

FIG. 25 illustrates an example of a mobile device using predistortion.

DETAILED DESCRIPTION

Various examples will now be described more fully with reference to theaccompanying drawings in which some examples are illustrated. In thefigures, the thicknesses of lines, layers and/or regions may beexaggerated for clarity.

Accordingly, while further examples are capable of various modificationsand alternative forms, some particular examples thereof are shown in thefigures and will subsequently be described in detail. However, thisdetailed description does not limit further examples to the particularforms described. Further examples may cover all modifications,equivalents, and alternatives falling within the scope of thedisclosure. Like numbers refer to like or similar elements throughoutthe description of the figures, which may be implemented identically orin modified form when compared to one another while providing for thesame or a similar functionality.

It will be understood that when an element is referred to as being“connected” or “coupled” to another element, the elements may bedirectly connected or coupled or via one or more intervening elements.If two elements A and B are combined using an “or”, this is to beunderstood to disclose all possible combinations, i.e. only A, only B aswell as A and B. An alternative wording for the same combinations is “atleast one of A and B”. The same applies for combinations of more than 2Elements.

The terminology used herein for the purpose of describing particularexamples is not intended to be limiting for further examples. Whenever asingular form such as “a,” “an” and “the” is used and using only asingle element is neither explicitly or implicitly defined as beingmandatory, further examples may also use plural elements to implementthe same functionality. Likewise, when a functionality is subsequentlydescribed as being implemented using multiple elements, further examplesmay implement the same functionality using a single element orprocessing entity. It will be further understood that the terms“comprises,” “comprising,” “includes” and/or “including,” when used,specify the presence of the stated features, integers, steps,operations, processes, acts, elements and/or components, but do notpreclude the presence or addition of one or more other features,integers, steps, operations, processes, acts, elements, componentsand/or any group thereof.

Unless otherwise defined, all terms (including technical and scientificterms) are used herein in their ordinary meaning of the art to which theexamples belong.

FIG. 1 schematically illustrates an example of a predistortion circuit100 for a wireless transmitter. The predistortion circuit 100 comprisesa signal input 110 configured to receive a baseband signal 102. Apredistorter 120 is configured to generate a predistorted basebandsignal 104 using the baseband signal 102 and a select of a firstpredistorter configuration 120 a or a second predistorter configuration120 b.

Predistortion can serve to linearize the output of a power amplifierused to amplify a wireless signal before transmission. Power amplifiersof wireless transmitters introduce nonlinearities depending on thebaseband signal to be amplified and on further operation characteristicsof the power amplifier. For example, if the power amplifier is operatedin saturation (in envelope tracking, ET), it tends to show morenonlinearities than in an operation mode where it is not operated insaturation (using average power tracking, APT). A predistortion circuitperforms a predistortion function on the baseband signal to alter thebaseband signal such that the nonlinearities of the power amplifier areanticipated and pre-compensated. If the predistortion function workedperfect, the baseband signal would be transformed into a pre-distortedbaseband signal such that an output of a power amplifier used to amplifya wireless transmit signal which is based on the baseband signal (e.g. aradio frequency signal generated by up-mixing the pre-distorted basebandsignal) is not deteriorated by the nonlinearities of the poweramplifier. In other words, the nonlinearities of the power amplifier areanticipated and inversely superimposed on the baseband signal, to resultwith an amplified radiofrequency signal having the desired signalcharacteristics and the desired signal quality. In other words, tolinearize the signal at the PA output, a predistortion circuit needs tocreate an IM spectrum that cancels the IM spectrum of the PA at itsoutput.

Examples of predistortion circuits allow choosing between at least twopredistorter configurations to generate the pre-distorted basebandsignal, while the predistorter functions corresponding to thepredistorter configurations may have different complexity.

In a first predistorter configuration 120 a, the predistortion circuit100 performs a first predistortion function and in the secondpredistorter configuration 120 b it performs a second predistortionfunction. The predistortion functions are used to modify the basebandsignal and to generate the predistorted baseband signal.

A complexity of a predistorter configuration or of the predistortionfunction performed within the configuration is for instance given by thenumber of multipliers and/or adders, or generally by the number ofcomputation nodes used within the presently chosen predistorterconfiguration to perform the predistortion function. A computation nodewithin a predistorter may be a hardware entity or a software portionthat performs a single fundamental computation like, e.g., an additionor a multiplication. Consequently, a more complex predistortion functionhas more computation nodes than a simpler one.

The complexity of a required predistortion function is, for example,driven by the transmit bandwidth and the memory contribution of thetransmit chain. For example, a higher number of multipliers may berequired for wideband linearization if the transmit bandwidth is higher.The same applies for memory contribution. If the transmit chain includesstrong memory effects (e.g. due to ET operation, particularly at theband edge where the filter skirt kicks in) then the complexity of thepredistortion function required may further increase leading to furtherincrease in the number of multipliers/computation nodes may furtherincrease.

In the event of complex models, predistortion might, on the other hand,have a significant impact on current consumption KPIs (key performanceindicators). For example, the number of multipliers may determine thecurrent consumption of the predistortion circuitry. In a conventionalapproach, the complexity of the predistorter configuration is driven bythe worst case waveform supported by the transmitter (e.g. maximumtransmit bandwidth, constellation density, peak-to-average power ratio),the worst case memory contribution of TX chain, required linearizationbandwidth, etc. so that a single predistorter configuration suites allpossible scenarios. However, for less complex waveforms supported by thewireless transmitter, a predistorter configuration with lower complexitymight be good enough to achieve the linearity performance or signalquality requirement as defined, e.g. by ACLR and EVM.

According to the example illustrated in FIG. 1, the predistorter isconfigured to use one of the at least two configurations that may allowto use a configuration consuming less power for a given operationcharacteristic while at the same time fulfilling all signal qualityrequirements. If a simpler configuration of the predistortion circuit issufficient for the given waveform to be amplified (depending on thepresent operation characteristics of the wireless transmit circuit) thensuch simpler configuration may be chosen so as to save or conserveenergy. The predistortion function nonetheless generates a predistortedbaseband signal fulfilling all signal quality requirements.

According to some examples, a first number of computation nodes isactive in the first predistorter configuration and a second number ofcomputation nodes are active in the second predistorter configuration.Assuming that the second predistorter configuration is more complex, thesecond number of computation nodes may be higher than the first number.If a simpler configuration of the predistorter is sufficient to amplifythe waveform of the baseband signal (e.g. first predistorterconfiguration may be a simpler configuration than the secondpredistorter configuration) then the predistorter circuit 120 may becapable of switching from the second predistorter configuration to thefirst predistorter configuration. Such flexibility to switch to thesimpler configuration of the predistorter (e.g., first predistorter 120a here) allows the energy to be saved since the first predistorterconfiguration requires less computation nodes to generate thepredistorted baseband signal.

Present mobile devices, already, support many different modulationschemes starting from relatively simple 3G voice signals up to morecomplex waveforms of, e.g., LTE-60 with 256 QAM. The introduction of 5GNR and the up-coming enhancements of the LTE standard will furtherincrease the spread between the “simplest”, least complex waveform andthe most complex waveform. Therefore, an unnecessary power consumptionof a conventional predistorter, which is designed to support the worstcase or most complex waveform is likely to increase with futurestandards. In a conventional approach, the current consumption of thepredistortion block or circuit would be unnecessarily high for a lot ofthe supported waveforms (e.g. LTE signals with low RB allocation) whichwould waste battery current and degrade the results in critical KPItests.

According to some examples, the predistortion circuit 100 comprises aconfiguration handling circuit configured to select the firstpredistorter configuration or the second predistorter configurationdepending on an operating characteristic of the wireless transmitter.The operating characteristics that may be used to select thepredistorter configuration can also depend on the mode of operation ofother elements within a wireless transmitter and/or on characteristicsof the baseband signal to be transmitted, as subsequently discussed withrespect to FIG. 2. FIG. 2 illustrates an example of a predistortioncircuit within a transmitter allowing to adapt the predistortercomplexity in response to predefined criteria. As a result, currentconsumption may be reduced, shorter calibration time may be achieved andmemory area to store digital pre-distortion (DPD) coefficients may besaved.

As generally illustrated in FIG. 2, the implementation of a DPD systemincludes several tasks like predistortion of a baseband signal in theforward path, monitoring and capturing the distorted signal by means ofan observation path and learning the predistortion function based on acomparison between reference data and data captured by the observationpath. In a predistortion system, coefficients corresponding to apresently chosen predistorter configuration (implementing acorresponding predistortion function) may so dynamically updated. Asillustrated in FIG. 2, a predistortion circuit 250 according to theexamples is located in the forward path and may thus implement thecorresponding techniques described below.

The observation path mainly comprises an observation block 210 which isconnected to an RF front-end subsystem 220. Observation block 210receives a portion of the distorted power amplifier (PA) signal 212 asits input signal 214. The distorted PA signal 212 may be captured bymeans of a coupler 222 that is part of the RF front-end subsystem 220.The input signal 214 is down-converted to baseband (BB) domain andfurther processed e.g. by running a time alignment with reference data,scaling, offset removal etc. within a signal conditioning block 230. Aprocessed input signal 216 which still includes the distortions of theTX signal is used as first input to DPD update block 240. Acorresponding reference signal 218 or corresponding reference data isthe second input to the DPD update block 240. The reference signalcorresponds to the baseband signal which was used to generate thedistorted PA signal 212. In further examples, the reference signal maybe derived from the output 252 of predistortion circuit 250. An examplefor such an alternate configuration is subsequently shown in FIG. 11.

The DPD update block 240 updates a predistortion function of thepresently used predistorter configuration (e.g. parameters used within aparticular configuration of predistortion circuit 250) by comparing asequence of the processed input signal 216 with a time aligned sequenceof the reference signal 218.

The configuration handling circuit 280 decides on the appropriatepredistorter configuration to be used within the predistortion circuit250 based on at least one of various operating characteristics of thewireless transmitter. A new predistorter configuration is transferred tothe predistortion block or circuit 250 and to the DPD update block 240.The predistortion circuit 250 may require the new predistorterconfiguration to select an appropriate or correct predistortion functionand apply the correct predistortion function to generate a predistortedbaseband signal. The predistorted baseband signal is provided to an RFsignal generation circuit 270, which upconverts the predistortedbaseband signal to a RF signal and provides the RF signal to PA 260 foramplification. The DPD update block 240 needs the new configuration todetermine and update the coefficients used by the new predistorterconfiguration within predistortion circuit 250. In other words,configuration handling circuit 280 is configured to select the firstpredistorter configuration or the second predistorter configurationdepending on an operating characteristic of the wireless transmitter.

According to some examples, the operating characteristic considered bythe configuration handling circuit 280 comprises at least one of anAverage Power Tracking Mode, an Envelope Tracking Mode, an output powerrange, a peak-to-average power ratio of the baseband signal, apeak-to-average power ratio of the input signal 214 (e.g. if the PA isin saturation or else introduces significant waveform clipping due toinsufficient power headroom for the instantaneous peaks of waveform, adifferent predistorter function might be beneficial), a modulationscheme used to generate the baseband signal, a matching condition of anantenna, a transmit bandwidth, a transmit band, a transmit frequencyrange within the transmit band, a number of transmit clusters in afrequency domain, a frequency separation between transmit clusters, abandwidth of each transmit cluster, and an acceptable spectral mask.While some examples may use only one of the above operatingcharacteristics, other examples may use an arbitrary combination thereofto conclude on the predistorter configuration to be used. For example,the individual criteria may receive different weights before combiningthem to conclude on the appropriate predistorter configuration.

FIG. 3 shows an example how the decision on the different configurationsmay be based on a bandwidth criterion (LTE-40/-60 orLTE-20/15/10/5/3/1.4) and on a mode criterion (APT/ET). Both graphsillustrate the transmit power on the y-axis and the used frequencyresources on the x-axis. The upper graph illustrates LTE 20configurations having a bandwidth of 20 MHz and below. The predistorterconfiguration depends on transmission bandwidth and DC/DC convertermode, which can be APT or ET. In the event of a comparatively lowtransmission power below a first threshold 303, the PA may be operatedin APT mode, since the overall power consumption is still moderate. Inthis setting and for the relatively uncomplex signal waveform of LTE20signals, predistortion may not be necessary at all, given that the PA inAPT mode may not show strong nonlinearity effects. For intermediatepowers 304 between the first threshold 303 and a second threshold 305,the PA may be required to operate in ET mode, resulting in strongernonlinearities. This may require the use of a predistorter model ofmoderate complexity, as for example a model having 3 coefficients asillustrated in column 308 of FIG. 3, upper graph. For high transmitpowers 306 above the third threshold 305, predistortion may also berequired with a complexity corresponding to the one of the ET mode atintermediate powers.

The lower graph illustrates LTE 40/60 configurations having basebandsignals with a bandwidth of 40 MHz and above. Like in the upper graph,the PA mode changes from APT for the lower power ranges 312, 314, and316 to ET modes for the higher power ranges 318 and 320. Due to thecomplex waveform of the high bandwidth LTE-40/60 signal, predistortionwith intermediate complexity may already be required for the APT mode.In the example illustrated in FIG. 3, the corresponding predistorterfunction has 8 parameters, as illustrated in column 322. The mostcomplex predistorter configuration is required for LTE40/60 signals athigh transmit powers 318 and 320, where the PA is operated in ET modeand where the predistorter function has 25 parameters. In the examplesof FIG. 3, the most complex predistorter configurations of transmitpowers 318 and 320 may consume more than three or four times the currentor power than that of the simplest configuration for intermediate powers304 in LTE 20 and below. Using examples of predistorter circuitsdescribed herein may allow to save the delta in current (for example,the difference between 25 parameters of power ranges 320 or 318 comparedwith the 3 parameters of power range 320) since they allow to switchbetween different predistorter configurations, each being sufficient forthe present setting of the wireless transmitter. With further increasingbandwidth of up-coming and future communication standards like, forexample, 5G, the potential saving is likely to significantly increase.

In further examples, the configuration handler 280 can change thepredistorter configuration that matches to a certain criterion furtherdepending on the feedback from the DPD update block 240, e.g. based on aresidual error after optimization, the condition of a matrix etc. Inthese examples, the configuration handler circuit 280 further comprisesan input interface configured to receive a feedback signal depending onan output of a power amplifier 260 of the transmitter, wherein theconfiguration handler circuit 280 is further configured to select thefirst predistorter configuration or the second predistorterconfiguration depending on the feedback signal.

While the examples of predistortion circuits are not limited to aspecific realization of the different predistortion functions thatcorrespond to the different predistorter configurations, one particularimplementation is subsequently described. In the following example,Volterra series based predistorter configurations are used, withoutlimiting the scope of further examples.

The following formula shows a generic base band (BB) representation of aVolterra series based predistorter.

${z(n)} = {{\sum\limits_{p = 1}^{K}{{z_{p}(n)}{with}{z_{p}(n)}}} = {\sum\limits_{k_{1} = 0}^{N}{\ldots{\sum\limits_{k_{p} = 0}^{N}{h_{k_{1},\ldots,k_{p}}^{(p)}{x\left( {n - k_{1}} \right)}{\prod\limits_{{u = 3},5,\ldots}^{p}{{x\left( {n - k_{u - 1}} \right)}{x^{*}\left( {n - k_{u}} \right)}}}}}}}}$

N is a memory depth, p is an order of the kernel and K is the maximumorder. It is again noted that the examples are not limited to aparticular polynomial representation. The polynomial representationabove is just an example for better understanding of the use ofdifferent predistorter configurations.

In this particular example, a single configuration corresponds to theactual set of used Volterra kernels. A Volterra kernel is herebycharacterized by the order of the polynomial p, by the memory depth Nand by the time indices.

EXAMPLES

x(n−k ₁)*|x(n−k ₁)|^(p-1)

x(n−k ₁)*|x(n−k ₂)|^(p-1)

x(n−k ₁)*x(n−k ₂)*x(n−k ₃)^(*)

Each kernel is weighted by a coefficient h(p)k₁ . . . kp. The sum of thekernels generates the predistorted baseband signal 252 that is used tolinearize the transmit chain by precompensating the nonlinearities of apower amplifier 260 which receives a radio frequency signal after itsgeneration within radio frequency generation block 270.

Changing a coefficient, however, does not generate a new configurationsince the coefficients are continuously updated by the DPD update block240 to achieve optimum predistortion by the presently selectedpredistorter configuration. A coefficient update from a first value thatis unequal to zero to a second value that is unequal to zero, therefore,changes the predistortion function but is not considered as a newconfiguration in the context of the examples.

Changing a coefficient from a non-zero value to zero or vice versa maybe interpreted as changing the configuration since a coefficient withvalue zero means that a kernel is not used.

In the context of the examples, a first configuration is different froma second configuration if a first set of active kernels (having a firstnumber of computation nodes) associated with a first configuration isdifferent from a second set of active kernels (having a second number ofcomputation nodes) associated with the second configuration. Activekernel means hereby that the kernel is used and contributes to thesignal that is generated by the predistortion circuit.

While examples of predistortion circuits can be used within allimplementations where predistorting a signal is desirable, the followingconsiderations detail an application in the field of wirelesscommunication. Some examples of digital predistortion (DPD) areimplemented to support LTE advanced or 5G modulation schemes. 5G NR (NewRadio) is a new communication standard with commercial deploymentexpected to start in 2020. 5G NR will feature higher bandwidth and morecomplex modulation schemes in the uplink than, for example, 4G. Forexample, mobile stations may need to support up to 200 MHz aggregatedbandwidth (2×100 MHz) in sub-6 GHz and more than 1 GHz in mmW range (>24GHz).

Baseline uplink modulation schemes that may be used in 5G NR areDFT-s-OFDM (which is similar to SC-FDMA used in LTE uplink) or CP-OFDM.

As already mentioned, the examples are not limited to modulation schemesaccording to LTE or 5G NR standard. Further examples are likewiseapplicable to any modulation scheme that imposes stringent linearityrequirements to the TX chain.

In summary, at least two challenges arise for mobile terminalssupporting 5G NR:

-   -   Higher channel bandwidth (up to 200 MHz) in the sub-6 GHz range,        more than 1 GHz expected in mmW range.    -   Signals with higher peak-to-power ratio (PAPR) to increase        throughput by using efficient modulation schemes like SC-FDMA        256QAM and OFDM.

More complex signals with higher PAPR and higher constellation density(e.g. 256QAM, 1024QAM) have more demanding requirements for thelinearity of the TX chain. More linearity requirements mean less AMAM-and AMPM conversion. This may conventionally be achieved by sufficientPA headroom preventing clipping of the modulation peaks and by increasedPA quiescent current flattening the AMPM response of the PA. Bothmeasures would, however, significantly increase the current consumptionof the PA. Digital predistortion (e.g. in the baseband domain) accordingto an example can be used to achieve the linearity requirements whilemitigating the increase of PA current consumption. By appropriatepredistortion of the PA input signal less PA headroom and PA quiescentcurrent are needed while maintaining sufficient linearity to, forexample, meet ACLR (Adjacent Channel Leakage Ratio) and EVM (ErrorVector Magnitude) targets.

FIG. 4 schematically illustrates a flowchart of an example of a methodfor generating a predistorted baseband signal for a wirelesstransmitter. The method comprises selection 410 to select a firstpredistorter configuration or a second predistorter configuration andpredistortion 420 for generating the predistorted baseband signal usinga baseband signal and the select predistorter configuration.

The above considerations rely on the following principles and furtherdecision criteria for selecting predistorter configurations may bedetermined based on the following considerations. An increase oftransmit bandwidth would likewise boost the complexity of thepredistorter since the linearity characteristic of the TX chain dependson the instantaneous RF frequency and how fast the envelope of modulatedRF signal changes. This is because the linearity characteristic dependson the IQ data stream which modulates the RF carrier. At higherbandwidths, AMAM- and AMPM response is not constant anymore and dependson the sequence of modulation symbols. This is also called memory effectand gets more pronounced at higher transmit bandwidths. There aremultiple potential effects that introduce a memory effect in a TX chain.The effects are partly different for ET (Envelope Tracking) and APT(Average Power Tracking) systems. Compared to APT operation, ETintroduces severe non-linear effects in transmit chain whereas thenon-linear effects partly depend on the RF frequency. As a result, thenon-linear characteristic can change within a few MHz causing adispersive TX channel. In a dispersive TX channel, the predistortioncharacteristic is strongly mapped to the absolute RF frequency. DPDcoefficients for a given predistortion function that are optimized for a1^(st) frequency range will show less linearity improvement in a 2^(nd)frequency range if the 2^(nd) frequency range is shifted by a few MHz ifthe channel is dispersive. To overcome the frequency dependence of theAMAM and AMPM response, memory predistortion is required.

Some relevant effects causing a dispersive channel in ET mode are:

-   -   ET delay dispersion over frequency    -   Too low tracker bandwidth cutting the envelope bandwidth and        introducing delay variations.    -   Too low VCC bandwidth in the PA module    -   PA load-pulling due to TX filter: This effect is more severe in        an ET system having a PA operating in compression, where it        exhibits higher load sensitivity. TX filters feature an input        impedance (=PA load impedance) that is highly frequency        dependent due to the resonators being used to form the filter        characteristic.

The ET delay dispersion and load-pulling due to TX filter are thedominating effects for a dispersive channel. While both effects alsooccur in the center of a TX band, they get more severe at the band edgeswith the transition from pass band to stop band.

At high transmit bandwidths (e.g. >400 MHz) APT operation may be a PAsupply scheme if an ET approach that can support >400 MHz withreasonable efficiency and system complexity may be difficult toimplement. In APT there are at least two dominant effects introducingmemory in a TX chain:

-   -   PA load-pulling due to the TX filter. The effect occurs also in        ET but is less pronounced in APT Operation where the PA operates        in its linear regime.    -   Limited PA bias bandwidth: This effect also occurs in an ET        system but is normally hidden by VCC bandwidth limitations in        the ET System. In APT systems, this effect is more dominant. A        well-designed PA bias network provides a low impedance from DC        up to BB modulation bandwidth. The low impedance is required to        avoid re-modulation effects, which introduce inter-modulation        errors in the spectrum of the PA output signal.

Signal deteriorations caused by limited PA bias bandwidth depend on theenvelope of the RF signal and on how fast the envelope changes.

PA load pulling effects depend on the instantaneous RF frequency and theimpedance response of the TX filter over the modulation bandwidth. Whenthe instantaneous modulation frequency is at a 1^(st) frequency, the PAis loaded by a 1^(st) impedance, when the instantaneous modulationfrequency is at a 2^(nd) frequency the PA is loaded by 2^(nd) impedance.Both impedances might be quite different especially if the separationbetween the instantaneous frequencies is large which may happen in caseof a signal with high transmit bandwidth. As a consequence, theresultant AMAM- and AMPM response at the 1^(st) frequency is differentfrom the response at the 2^(nd) frequency.

To linearize a PA output in such circumstances, the predistortioncircuit needs to create an inter modulation (IM) spectrum that cancelsthe IM spectrum of the PA at its output. The demands for thepredistortion circuit are exemplarily described assuming a transmitbandwidth of 1 GHz. The bandwidth of the IM3 spectrum (→x{circumflexover ( )}3) is then 3 GHz, the bandwidth of the IM5 spectrum(→x{circumflex over ( )}5) is 5 GHz. The predistorter needs to generatethe IM3- and IM5 spectrum in BB domain to achieve the desired goal oflinearization of the PA output. If, for example, both IM3 and IM5spectrums shall be cancelled, the required bandwidth in BB domain is+/−2.5 GHz. As a result, the computation nodes (e.g. multipliers) thatare used to generate the predistortion signal need to operate at asample rate of 5 GHz to preventing overlapping (aliasing) of the IM5spectrum with its replicas in the frequency domain.

However, computation nodes or multipliers operating at 5 GHz may drawsignificant current.

The example of a predistortion circuit 500 illustrated in FIG. 5 allowsto deploy memory predistortion in transmit systems featuring a hightransmit bandwidth while reducing current consumption and designcomplexity of the predistorter circuit 500. FIG. 5 illustrates anexample where the different predistorter configurations may distinguishby the rate at which groups of computation nodes within the differentconfigurations are operated at.

In the example of FIG. 5, the predistortion circuit 500 is separatedinto 3 parts or sub-circuits 510, 520, and 530 (also referred to asblocks), which may contribute to a single predistorter configuration orwhich may contribute to different predistorter configurations inarbitrary combinations of the sub-circuits 510, 520, and 530. Thecomputation nodes of each sub-circuit 510, 520, and 530 operate atdifferent sample rates. In particular, a first number of computationnodes are configured to operate at a first rate and a second number ofcomputation nodes are configured to operate at a second rate, the secondrate being higher than the first rate. Further examples may also have 2sub-circuits that are selectively or jointly activated for apredistorter configuration. More than three sub-circuits may also beappropriate depending on the predistortion function to be implementedand on the order of IM spectra to be considered. Further examples arenot limited to a specific number of blocks or to specific mathematicaloperations executed by a block. The total predistortion functionperformed by the predistortion circuit 500 is broken down in partialfunctions with different bandwidth requirements and the partialfunctions are mapped to sub-circuits that run at different appropriaterates.

If all sub-circuits are used within a predistorter configuration, eachsub-circuit generates a partial signal 512, 514, and 516 as a part ofthe predistortion signal 518. The partial signals 512, 514, 516 arecombined at the output of the sub-circuits 510, 520, 530 after applyinga rate matching to equalize different sampling rates. For example, asample rate converter may be configured to match an output of the firstnumber of computation nodes to the second rate to generate a ratematched output of the first number of computation nodes. Further, acombination circuit may be configured to combine an output of the secondnumber of computation nodes and the rate matched output of the firstnumber of computation nodes. In FIG. 5 it is assumed that sub-circuit510 (#1) comprises all linear terms that do not increase the bandwidthof the input signal, sub-circuit 520 (#2) comprises all 3^(rd) orderkernels generating a signal with 3 times the bandwidth of the inputsignal, and sub-circuit 530 #(3) comprises all 5^(th) order kernelsgenerating a signal with 5 times the bandwidth of the input signal.Therefore, sub-circuit 520 operates at 3 times the sampling rate ofsub-circuit 510, and sub-circuit 530 at 5 times the sampling ofsub-circuit 510 to avoid aliasing.

The highest sampling rate and the highest current consumption is,therefore, only spent for 5^(th) order kernels where the highestsampling rate is required to avoid aliasing. The same applies to the3^(rd) order kernels. The example of a predistortion circuit 500 ensuresthat the computation nodes of the predistortion circuit 500 only operateat a sampling rate that is required to prevent aliasing. The example ofa predistortion circuit 500 helps to save energy as compared to a singlerate implementation where all computation nodes (multipliers) would runat highest rate. The example of a predistortion circuit 500 also saveenergy as compared to a polyphase implementation that would increase thenumber of multipliers operating at a lower sampling rate, which mightreduce some design challenges at high sampling rates but would not helpto decrease current consumption.

As compared to alternative approaches, the current consumption ofpredistorter circuit 500 is significantly reduced by running thecomputation nodes (e.g. multipliers) at different rates depending on,e.g., the signal bandwidth.

FIG. 7 illustrates the deterioration the spectrum of the predistortedbaseband signal might suffer from due to aliasing if an insufficientrate would be chosen for the operation of the predistortion circuit. InFIG. 7 it is assumed that Intermodulation distortions of 3^(rd) ordershall be considered by the predistortion function. The main spectrum 710of the predistorted baseband signal overlaps with the first replica 712of the predistorted baseband signal so as to deteriorate the quality ofthe predistorted baseband signal if the sample rate of the computationnodes was chosen too low. In using a predistorter configurationaccording to an example of a predistortion circuit, the sampling ratemay be chosen sufficiently high for every IM order to be consideredwithout increasing the current consumption more than necessary toachieve this goal. In other words, FIG. 7 illustrates the harmfulinterference of the IM spectrum caused by a too low sampling rate. InFIG. 3, the overlapping of the IM3 spectrum will degrade thelinearization result since the overlap adds unwanted IM contributions inthe spectrum of the predistorter circuit.

FIG. 6 shows a further example of a predistortion circuit 600. Firstorder computations are performed at the lowest rate within the firstsub-circuit 610 (#1), second order computations are performed at anintermediate rate within the second sub-circuit 620 (#2) and third ordercomputations are performed at the highest rate within the thirdsub-circuit 630 (#3). Other than in the example of FIG. 5, thesub-circuits 610, 620, 630 are serially connected and comprise samplerate converters 612, 622, and 632 to adjust the sample rates between thesubsequent sub-circuits of the serially connected chain of sub-circuits.

In a further example it is implemented to use intermediate resultsgenerated in a first block in at least a second block after appropriaterate matching to mitigate the total number of multipliers. This isillustrated by the vertical connections between the sub-circuits 510,520, and 530 in FIG. 5.

In summary, digital predistortion for ultra-high bandwidth signals inmobile terminals may be heavily constrained by the battery currentconsumption. In mobile terminals the current consumption is key forcompelling customer experience and for solving critical heat dissipationissues that occur in a small form factor device. For example, apredistorter configuration with moderate complexity may already require30 to 60 multipliers for very high bandwidth signals, which may draw acouple of 100 mA of the predistortion circuitry (in addition to the PAcurrent), depending on technology and digital implementation.

The increase of transmit bandwidth being introduced by 5G NR will createnew challenges for mobile terminals. The increase of transmit bandwidthis likely to be faster than the deployment of new process nodes thatintentionally improve the baseline current year by year (e.g. 28 nm, 16nm, 10 nm). As a non-favorable consequence the current consumption ofconventional digital predistorters would heavily increase despite of newprocess nodes. There is a strong need to introduce new methods andevolved digital circuitries so that current consumption does less scalewith the increasing transmit bandwidth. Otherwise the excessive currentconsumption of digital part at ultra-high bandwidth will prevent ordelay its deployment. The increase of current consumption can bemitigated by the examples described herein.

In summary, examples of the methods and predistortion circuits describedpreviously run the calculations that are required to generate thepredistorted signal on at least two sampling rates whereas the samplingrate depends on the bandwidth of the signal after applying thepredistortion specific mathematical operations. An example of apredistortion circuit comprises at least two sub-circuits whereas eachsub-circuit generates a partial signal of the predistorted signal andwhereas a sampling rate of a first sub-circuit is different from asampling rate of a second sub-circuit.

Examples of predistortion circuits could help to deploy predistortionfor signals with ultra-high bandwidth that for instance will occur in 5GmmW applications. They mitigate the increase of current consumption anddesign complexity due to the upcoming boost of transmission bandwidth.Examples of predistortion circuits allow to provide mobile handsets insmall form factor and with leading edge battery current consumption.

For example, in a 5G NR mobile terminal, predistortion circuits areexpected to have a significant impact on battery current (4 operationtime) and dissipated power/heating (4 device size and form factor).

A particular example of a predistortion function suitable to implement apredistortion circuit according to FIG. 5 or FIG. 6 is a Volterra seriesbased function. The following formula shows the generic BBrepresentation of a Volterra based predistorter.

${z(n)} = {{\sum\limits_{p = 1}^{K}{{z_{p}(n)}{with}{z_{p}(n)}}} = {\sum\limits_{k_{1} = 0}^{N}{\ldots{\sum\limits_{k_{p} = 0}^{N}{h_{k_{1},\ldots,k_{p}}^{(p)}{x\left( {n - k_{1}} \right)}{\prod\limits_{{u = 3},5,\ldots}^{p}{{x\left( {n - k_{u - 1}} \right)}{x^{*}\left( {n - k_{u}} \right)}}}}}}}}$

N is the memory depth, p is the order of the kernel and K is the maximumorder. However, further examples are not limited to this specificpolynomial representation. The polynomial representation is just givento better illustrate the meaning of the x{circumflex over ( )}3,x{circumflex over ( )}5, x{circumflex over ( )}7 terms in a widercontext.

While the previous examples address the forward path of a systememploying predistortion, further examples address the observation pathand a control circuit used to update the predistortion parameters whichare used within a predistortion circuit to generate a predistortedbaseband signal.

An example of a control circuit 800 for a predistortion circuit isillustrated in FIG. 8. The control circuit 800 for a predistortioncircuit comprises a feedback signal input 810 configured to receive afeedback signal 812 related to an output of a power amplifier as well asbandwidth limitation circuitry 820 configured to limit a bandwidth ofthe feedback signal 812 to derive a bandlimited feedback signal 822. Aparameter handling circuit 830 is configured to update predistortionparameters used within the predistortion circuit based on thebandlimited feedback signal 822. Optionally, further examples may alsocomprise an output interface 840 to output the updated predistortionparameters. The limitation of the bandwidth may be performed by anarbitrary filter circuit in a digital or an analog domain. For example,a FIR filter may be used in the digital domain.

Using a bandlimited feedback signal 822 to update the predistortionparameters used within predistortion circuit may significantly decreasepower consumption of the control circuit 800 and of further componentswithin the feedback or observation path since power consuming componentscapable of operating at high bandwidths may be avoided.

Conventional implementations, however, use a continuous and broadspectrum for predistortion learning and to calculate the updatedpredistortion parameters. Predistortion learning describes the processof generating a predistortion function that is intended to compensatenonlinear effects introduced by the analog transmit system. Forpredistortion learning, an observation path is required that captures aportion of the distorted RF output signal of the transmit system (as,e.g., generated by a power amplifier) and down-converts it to BB domainfor further processing, as for example illustrated in FIG. 9. For theexample, for the future communication standards already elaborated onbefore, energy savings of an example illustrated in FIG. 8 may besignificant as compared to a conventional approach. Assuming a signalbandwidth of 1 GHz, the IM spectrum may exhibit a bandwidth of 5 GHz(covering IM3 and IM5 products). Thus, the observation path of aconventional solution would have to be flat, both in terms of delay andamplitude, over a bandwidth of 5 GHz in the RF domain and, for example,an ADC sampling rate of 5 GHz would be required to meet Nyquist'scriteria. A 5 GHz ADC, however, is complex and draws significantcurrent.

The present examples of control circuits, instead, offer a compellingsolution how to deploy memory predistortion in transmit systemsfeaturing a high transmit bandwidth (e.g. 5G NR) while reducing currentconsumption and design complexity of the predistortion implementation.The subsequently illustrated examples address the observation of thesignal output by a power amplifier and introduce a concept to reduce thecomplexity of the observation path.

FIG. 9 illustrates an example of a control circuit 940 within atransmitter circuit. The transmitter circuit comprises a predistortioncircuit 910 configured to receive a baseband signal 912 and to generatea predistorted baseband signal 922. To generate the predistortedbaseband signal 922, the predistortion circuit uses a selectpredistorter model. The predistortion function of the predistorter modeluses predistortion parameters which are dynamically updated to optimizethe result of the predistortion. The predistortion parameters arecontinuously updated based on feedback from an amplified radio frequencysignal 932 by means of a control circuit 940. An upmixer 920 generatesthe radio frequency signal using the predistorted baseband signal 922and a local oscillator signal. A power amplifier 930 is coupled to anoutput of the upmixer 920 and generates an amplified radio frequencysignal 932.

The control circuit 940 for the predistortion circuit 910 comprises afeedback signal input 941 configured to receive a feedback signal 934related to an output of a power amplifier 930. According to someexamples, feedback signal 934 is a copy of the amplified radio frequencysignal 932 having low power. Bandwidth limitation circuitry 942 isconfigured to limit a bandwidth of the feedback signal 934 to derive abandlimited feedback signal 936. The control circuit 940 furthercomprises a baseband signal input 943 configured to receive the basebandsignal or reference signal 912. Further bandwidth limitation circuitry944 is configured to limit a bandwidth of the baseband signal 912 toderive a bandlimited baseband signal 914. A parameter handling circuit946 is configured to update the predistortion parameters used within thepredistortion circuit 910 based on the bandlimited feedback signal 936and on the bandlimited baseband signal 914. In the particular example ofFIG. 9, the bandlimited feedback signal 936 is further digitized by meanof an ADC 945. Further signal shaping is performed by means of a signalconditioning circuit 947 and a sampling circuit 949 to be able todirectly compare the bandlimited feedback signal 936 (as output fromsampling circuit 949) and the bandlimited baseband signal 914. Forexample, signal conditioning circuit 947 and sampling circuit 949 mayperform sample rate conversion and/or time alignment so thatcorresponding samples of the bandlimited feedback signal 936 andbandlimited baseband signal 914 can be directly compared by means of acomparator circuit 946 a which is part of parameter handling circuit940. Based on the comparison, update circuit 946 b within parameterhandling circuit 946 calculates and updates the predistortion parameterswhich are then communicated to predistortion circuit 910 to close theloop of control and to optimize the linearization achieved by the selectpredistorter model.

In other words, the transmitter circuit/communication system of FIG. 9comprises a forward path where digital predistortion is used to improvethe linearity of a transmit signal, an observation path that is coupledto the transmit path so that a coupled signal (feedback signal 934)includes the non-linear distortions of the transmit path, a comparatorblock that compares the coupled signal with a reference signal, and anupdate block that changes the predistortion function in the forward pathbased on a result of the comparator block.

In summary, an update of the predistortion parameters is performed basedon a comparison between the feedback signal and a reference signal atdiscrete frequencies points within the transmission spectrum. Anobservation bandwidth for the feedback signal and the reference signalmay be chosen such that there is no overlapping of adjacent observationchannels. The reference signal may be derived from the baseband signalbefore predistortion as illustrated in FIG. 9. Further examples maylikewise derive the reference signal from the predistorted basebandsignal at an output of the predistortion circuit.

FIG. 10 illustrates the idea of using discrete sample intervals in thefrequency domain instead of using the full spectrum to determine thepredistortion parameters. Both signals, reference and feedback signal,are sampled at identical frequency offsets with identical measurementbandwidth. FIG. 10 schematically illustrates the spectrum inside thetransmission bandwidth 1010 and of neighboring frequency portions 1020 aand 1020 b. The width of the neighboring frequency portions 1020 a and1020 b to be monitored depends on the order of the IM distortions to beconsidered.

FIG. 10 illustrates a particular example having 13 observation channels1030 a to 1030 m of limited bandwidth. The predistortion parameters maybe updated independently for each of the observation channels 1030 a to1030 m. According to some examples, the observation channels 1030 a to1030 m are processed subsequently in a predetermined order to considercontributions from the entire spectrum for updating the predistortionparameters. The spacing between neighboring observation channels doesnot need to be equidistant. Depending on the response of theintermodulation spectrum it may also be beneficial to select closerseparation at a first frequency offset, for example where the IMspectrum change over frequency is more pronounced compared to the changeat a second frequency offset. As illustrated in FIG. 10, the spectrum isdivided into chunks 1030 a to 1030 m for both the reference spectrum ofthe reference signal and the coupled spectrum of the feedback signal andthe predistortion function (the predistortion parameters) is updatedbased on a comparison of the chunks.

According to some examples, the compared chunks of the reference signaland the feedback signal have the same bandwidth. However, neighboringchunks may have different bandwidths.

As illustrated in FIG. 10, examples use narrowband signals fordetermination of the predistortion parameters. According to someexamples, the measurement bandwidth is a few megahertz, e.g. 1 MHz, 3MHz. 5 MHz, 10 MHz. Using the bandlimited signals may greatly reducecomplexity and current consumption. Further, the determination of thepredistortion parameters (DPD coefficients) can be done at lower rate insome examples, which depends on the measurement bandwidth, and, e.g., onNyquist's formula. However, the predistortion circuit or block itselfwhich creates the predistorted baseband signal may run at full rate toavoid aliasing of the IM spectra generated by the predistorter circuit.

FIG. 11 shows an example as to how the corresponding frequency chunksmay be generated. As opposed to the example of FIG. 9, the referencesignal is generated from the predistorted baseband signal at an outputof predistortion circuit 1110. Like in FIG. 9, it is desired to reducethe bandwidth of the observation path by narrowband sampling a feedbacksignal and a reference signal in the frequency domain. As a result, thebandwidth of the components within the observation path does not need tobe high enough to capture the complete feedback signal, resulting in adecrease of complexity and power consumption. Similar to FIG. 9, FIG. 11illustrates an upmixer 1120 to generate a radio frequency signal using alocal oscillator signal 1121. A power amplifier 1130 amplifies the RFsignal. A control circuit 1140 comprises bandwidth limitation circuitry1142 to limit a bandwidth of the feedback signal to derive a bandlimitedfeedback signal 1146. In the example of FIG. 11, the bandwidthlimitation circuitry 1142 comprises an adjustable mixer 1143 configuredto downmix the feedback signal 1141 using a feedback oscillator signal1144 to generate a downmixed feedback signal 1145, The feedbackoscillator signal 1144 has the frequency of the local oscillator signal1121 plus an observation frequency offset Δf.

A band pass filter circuit 1147 is configured to band filter thedownmixed feedback signal 1145 to generate the bandlimited feedbacksignal 1146. Prior to filtering, the downmixed feedback signal 1145 iddigitized by ADC 1149. In summary, the bandwidth limited feedback signalis generated by first downmixing the RF signal with a frequency of thefeedback oscillator signal and by subsequent bandpass-filtering thesignal to arrive at an observation frequency chunk centered at thefrequency of the feedback oscillator signal with a bandwidth as given bythe bandpass filter.

The control circuit 1140 further comprises a reference signal generationcircuit 1150 comprising a reference signal input configured to receivethe predistorted baseband signal 1111 form an output of thepredistortion circuit 1110. The reference signal generation circuit 1150comprises further bandwidth limitation circuitry configured to limit abandwidth of the reference signal to derive a bandlimited referencesignal. The further bandwidth limitation circuitry comprises a frequencyshifter 1151 configured to shift the predistorted baseband signal by thefrequency offset and a further band pass filter circuit 1153 configuredto bandpass filter the shifted predistorted baseband signal 1111 togenerate a bandlimited predistorted signal 1114. The bandlimitedpredistorted signal 1114 and the bandlimited reference signal are socreated as corresponding frequency chunks within the spectrum so that anoptimization circuit 1155 may directly compare the two signals to deriveupdated predistortion parameters. In other words, further bandwidthlimitation circuitry 1150 comprises a frequency shifter 1151 configuredto shift the reference signal by the inverse of the observationfrequency offset.

In summary, to realize appropriate frequency domain sampling, the LOfrequency of the mixer in the observation path is swept and said LOfrequency is typically set to the mid frequency of the frequency chunks.The LO frequency sweeps from chunk to chunk, the signal is downconverted, filtered (so that reference signal/predistorted basebandsignal and coupled feedback signal experience the same filtering) andfed to optimization circuit 1155 (optimizer block).

When sweeping the LO frequency, LO phase at a new frequency might beunknown in some circumstances. An unknown LO phase would preventlearning of the predistorter coefficients since the optimization circuit1155 could not distinguish if an observed phase shift is caused by PA orby LO frequency change. Some examples optionally include switch 1170which can be configured to connect an input signal from before the PA1130 to the mixer 1143. In a first measurement, the input signal maythen be connected to the mixer 1143. Since the input phase does then notinclude the PA phase distortions the signal can be used to determine theLO phase without contributions from PA 1130. In a second measurement,the output signal of the PA feedback signal 1145 may then be measured asan input to the optimizer circuit 1155. In other words, control circuit1140 may further comprises a phase control input coupled to an output ofthe upmixer to receive a phase control signal, wherein the controlcircuit 1140 is further configured to adjust a phase of the feedbacksignal based on the phase control signal. FIG. 12 again schematicallyillustrates the processing performed in the example of FIG. 11 to builda system of equations intended to determine the predistortioncoefficients for a select predistorter configuration.

During the optimization process a system of linear equations is solved,e.g. by means of the least square (LS) algorithm. FIG. 15 illustrates anexample for such a system of linear equations for a predistortionfunction based on a Volterra Series. As illustrated in FIG. 15, inMatrix A, reference data is used to calculate polynomials, each elementreflecting one Volterra kernel. The number of columns, K, equals thenumber of coefficients, while the number of rows, M, equals the numberof data points. Vector ‘h’ includes model coefficients that shall bedetermined, where K is the number of coefficients, and Vector ‘b’represents measured data, where M is the number of data points.

The lines of the matrix A are derived from different chunks orobservation channels, at least one line for each observation channel.For an observation channel, data may be accumulated as illustrated inFIG. 12. Reference signal 1210 and feedback signal 1220 may be collectedand bandwidth limited by means of filters 1230 and 1240, respectively.In order to be able to use the samples of both signals to calculate thepredistortion parameters, further matching of the signals samples may beperformed within optimization circuit 1255. For example, rate matchingcircuits 1262 and 1264 may be used to align the sample rates.Optionally, a phase calibration may be performed for the feedback signalas illustrated above. To this end, an optional phase correction circuit1266 for the local oscillator signal 1144 may be present as well as asubsequent time alignment circuit 1268. After compensating ambiguitiesand proper time alignment, the samples of the reference signal 1210 andof the feedback signal 1220 may be used as an input to the linearequations illustrated in FIG. 15.

Each line of Matrix A includes a reference data point that ispredistorted by the predistortion function (from the bandlimitedpredistorted baseband signal). Each predistorted data point of vector bis compared with a point of the coupled data/the bandlimited feedbacksignal of vector h. This is done for a large number of data points (e.g.a few thousands) and the predistortion coefficients h_(k) are determinedsuch so that they meet a certain metric e.g. least square error.

According to some examples, the optimization circuit 1155 may run at alow rate if low pass filtered data is used.

FIG. 13 again illustrates as to how the bandwidth limitation circuitryand the further bandwidth limitation circuitry of FIG. 11 may cooperateto guarantee that corresponding frequency chunks of the predistortedbaseband signal and the feedback signal are used by means ofoptimization circuit 1155.

FIG. 14 is an example how the further examples may be used in mmWapplications. In mmW designs, it may be attractive to have a RF head1410 that includes the critical analog functions for transmit, receiveand antenna beamforming. Since a communication device will have multipleheads for MIMO purposes, the RF signals and the LO signals are providedby special coax cables. As a consequence, the observation path may bemultiplexed within a transmitter circuit 1430.

In summary, the previous examples may enable to practically usepredistortion for signals with ultra-high bandwidth that for instancewill occur in 5G mmW applications. It will mitigate the increase ofcurrent consumption and design complexity due to the upcoming boost oftransmission bandwidth. The examples address the observation path andintroduce methods reducing the complexity of the observation path byusing multiple narrowband signals instead of a single wideband signalwhich may provide at least for the following:

-   -   A lower sample rate of the ADC (factor of 100 or more depending        on signal bandwidth).    -   Relaxed requirements for the amplitude- and frequency response        of the observation path due to narrowband signals→less        compensation efforts in digital domain to equalize the        observation path and less critical RF analog design.

Relaxed requirements for delay error between coupled signal andreference signal. As of today a signal with a bandwidth of 60 MHz cantolerate 1 . . . 3 ns delay error. For 1 GHz bandwidth the tolerableerror will be <<1 ns which would make it difficult to find a practicalimplementation for full bandwidth monitoring.

Further, cellular data transmission standards my not only rise bandwidthbut also output power. In conventional implementations, a mobileterminal might transmit with a maximum of 23 dBm. Up-coming power class2 defines 26 dBm output power for all TDD bands, power class 1 defines31 dBm output power for low band Band14. Due to high bandwidth, highchannel frequencies, power saving envelope tracking techniques andhigher output power, the time dependent nonlinearities (i.e. memoryeffects) of the analogue circuitries become more and more visible. Sincenonlinearities create higher frequencies (harmonics), the computationsrelating to a predistortion model have to be calculated at high samplingrates. Many multiplications due to complex mathematical models (VolterraSeries), high wordlengths (due to exponentiation for the harmonics) andhigh sampling frequency (due to frequency expansion of harmonics) mayresult with consumption of a lot of current.

FIG. 16 illustrates an example of an apparatus 1600 for predistorting abaseband signal 1601 which may allow to (significantly) reduce thecurrent consumption.

Apparatus 1600 comprises a predistortion circuit 1610 configured todetermine samples of the baseband signal 1601 at a first sample rate.For example, the first sample rate may be equal to or higher than asample rate of the baseband signal 1601. Further, predistortion circuit1610 is configured to calculate and output samples of a predistortedbaseband signal 1602 at a second sample rate based on predistortionparameters (e.g. based on a Volterra series) and the samples of thebaseband signal 1601. The second sample rate is lower than the firstsample rate. That is, predistortion circuit 1610 uses an output samplerate for calculating the predistorted baseband signal 1602 that is lowerthan the input sample rate of the baseband signal 1601.

Apparatus 1600 further comprises an upsampling filter 1620 configured tocalculate samples of the predistorted baseband signal 1602 at the firstsample rate based on the samples of the predistorted baseband signal1602 at the second sample rate.

The higher first sample rate may allow to track all higher orderexponential harmonics in the baseband signal 1601 so that the course ofthe baseband signal 1601 is known with sufficient precision. On theother hand, the lower second sample rate may allow to save current sincethe predistortion is done at this lower sample rate. The upsamplingfilter then restores the input sampling rate of the predistortioncircuit 1610. However, since upsampling filters use fixed coefficientsand since no exponential order occurs, the mere upsampling calculationis much simpler than the predistortion at higher sample rate.Accordingly, current may be saved and a reasonable attenuation of theharmonics may be achieved.

This may become more evident from the following non-limiting numericalexamples, which are given in conjunction with FIGS. 17 to 20.

FIG. 17 illustrates one period length from —7E to 7E of an analoguesignal 1710 according to the expression |Σ_(n=1 . . . 5) exp (It)/n|.Signal 1710 consists of a carrier plus its attenuated 2^(nd) 3^(rd),4^(th) and 5^(th) harmonics. Further, FIG. 17 illustrates digitalsamples 1700, . . . , 1709 of the signal. Signal 1710 is ten timesoversampled in the example of FIG. 17.

FIG. 18 illustrates the same signal 1710. However, in the example ofFIG. 18, signal 1710 is only five time oversampled—as indicated bydigital samples 1801, . . . , 1805.

Assuming that signal 1710 represents the course of a baseband signalthat is to be predistorted, it is evident from FIGS. 17 and 18 that apredistortion circuit/algorithm (e.g. for Memory Digital PreDistortion,MDPD) can hardly estimate/follow the 5^(th) exponential order harmonicat the low sample rate. In the example of FIG. 18, the digital sample1803 has an amplitude value of 1.8, while a subsequent power amplifieractually receives the analogue input signal with a maximum amplitudevalue of 2.3. Accordingly, a predistortion circuit/algorithm for MDPDwill try with several weighted exponential functions to predistort thedigital sample value 1.8, while the correct digital sample should be2.3. With ten time oversampling as illustrated in FIG. 17, the error ismuch smaller (digital sample 1705 has an amplitude value of approx. 2.3)than for five time oversampling.

An example of a baseband signal 1910 and an accordingly predistortedbaseband signal 1920 is illustrated in FIG. 19. Compared to basebandsignal 1910, the predistorted baseband signal 1920 comprises additionalsignal components relating to the 3^(rd) and 5^(th) order harmonics (3rdand 5^(th) order intermodulation distortions). Compared to the wantedsignal components around the carrier frequency (around 0 MHz frequencydeviation), the higher order exponential harmonics are attenuated.

Predistortion circuit 1610 of apparatus 1600 allows to track to higherorder exponential harmonics since the first sample rate (i.e. the inputsample rate) is high enough. Further, the second sample rate (i.e. theoutput sample rate) is low enough to save current and to achievereasonable attenuation of the higher order exponential harmonics. Inother words, apparatus 1600 may be understood as a novel downsamplingMDPD approach. In a second step, (higher order) upsampling filter 1620allows to restore the original MDPD input sample rate. Since theupsampling filter 1620 uses fixed coefficients and since no exponentialorder occurs, it is much simpler than the MDPD calculation (e.g.Volterra series). Apparatus 1600 may, hence, allow to save current.

Although the downsampling of predistortion circuit 1610 may lead to analias of the 5^(th) order harmonics in the predistorted baseband signal,the adaption algorithm of upsampling filter 1620, which receives thedownsampled signal as input, will optimize the addition of the aliasedsignal to the original signal inherently.

The effect of predistortion as described above in conjunction with FIGS.16 to 19 is illustrated in FIG. 20. FIG. 20 illustrates the spectra oftwo LTE 20 signals 2010 and 2020. Signal 2010 is generated based on abaseband signal a without predistortion, whereas signal 2020 isgenerated based on the same baseband signal using apparatus 1600 forMDPD.

The input signal for MDPD is a LTE carrier aggregation signal with ±19MHz bandwidth. A Volterra based 5^(th) order MDPD is used forpredistortion. The input sample rate of the predistortion circuit (i.e.the first sample rate) is 184 MHz, whereas the output sample rate of thepredistortion circuit is 92 MHz. This sampling frequency yields to anMDPD edge of 92/2 MHz=46 MHz. By comparing signals 2010 and 2020, it isevident, that MDPD according to the proposed concept provides signalattenuation up to the Nyquist rate (i.e. half of the sample rate, whichis 46 MHz in the example of FIG. 20). Accordingly, ACLR may be reduced.

In some examples, it may however be beneficial to not reduce the samplerate. This is exemplarily illustrated in FIGS. 21 and 22. FIG. 21illustrates the spectra of two LTE 20 signals 2110 and 2120. Signal 2110is generated based on a baseband signal a without predistortion, whereassignal 2120 is generated based on the same baseband signal usingdownsampling MDPD. The input signal for MDPD is again a LTE carrieraggregation signal with ±19 MHz bandwidth. A Volterra based 5^(th) orderMDPD is used for predistortion. As can be seen from FIG. 21, the LTEcarrier aggregation signal comprises two narrowband allocated spectrathat are widely separated from each other (i.e. a horn spectrum). Secondaliases for the 3^(rd) and 5^(th) order harmonics are created in the LTE20 signal 2120 due to the downsampling MDPD.

By performing the MDPD without downsampling, the creation of aliases maybe avoided and the 3^(rd) as well as the 5^(th) order harmonics may bereduced. This is illustrated in FIG. 22 illustrating the spectra of twoLTE 20 signals 2210 and 2220. Signal 2210 substantially corresponds tosignal 2110 illustrated in FIG. 21. Signal 2220 is generated based onthe same baseband signal as signal 2210 using MDPD without downsampling(i.e. normal MDPD).

Since downsampling MDPD may be adverse for certain rare spectra,apparatus 1600 may be adapted accordingly. In particular, thepredistortion circuit 1610 may be further configured to receiveinformation about a spectral allocation of the baseband signal's data ina frequency spectrum (i.e. the shape of the resulting signal spectrum;e.g. indicated by the allocated resource blocks). If the spectralallocation satisfies a first decision criterion (minimum/maximumbandwidth of allocated spectra or minimum/maximum distance betweenallocated spectra; e.g. spectrum as illustrated in FIG. 20), thepredistortion circuit 1610 may be further configured to calculate thesamples of the predistorted baseband signal at the second sample rate(i.e. the input sample rate). If the spectral allocation satisfies asecond decision criterion (e.g. spectrum as illustrated in FIG. 22), thepredistortion circuit 1610 may be further configured to calculate thesamples of the predistorted baseband signal at the first sample rate(i.e. the lower output sample rate). Accordingly, the upsampling filter1620 may be deactivated, if the spectral allocation satisfies the seconddecision criterion.

In other words, apparatus 1600 may support multirate DPD. According tothe known transmit signal, the circuitry or the algorithm forpredistortion may differ. For example, it may run with N multipliers indecimation mode or it may run with N/n multipliers for each polyphaserat a n-times higher sample rate. Alternatively, an improved upsamplingfilter which better suppresses the alias components might be used.Further alternative, MDPD might be switched off and the PA might beoperated in the more linear average power tracking mode.

An example of a wireless transceiver 2300 using downsamplingpredistortion is illustrated in FIG. 23. Baseband circuitry 2330provides a baseband signal 2301 to predistortion circuit 2310.Predistortion circuit 2310 samples baseband signal 2301 at its samplerate and outputs a predistorted baseband signal 2302 at a lower samplerate. Upsampling filter 2320 upsamples the predistorted baseband signal2302 to the original sample rate of baseband signal 2301.

The predistorted baseband signal 2302 is then upmixed to radio frequencyusing upmixer 2340 and further converted to an analog representation byADC 2350. The analog predistorted baseband signal is amplified by PA2360 and provided to antenna 2390 via duplexer 2380.

As indicated in FIG. 23, the supply voltage V_(cc) for PA 2360 isprovided by an envelope tracking circuitry 2370 (comprising an envelopetracking path for determining the envelope of the predistorted basebandsignal 2302, an ADC for digitalization and a DC-to-DC converter forproviding V_(cc) based on the digitized envelope information).

The downsampling MPDP used in wireless transceiver 2300 may allow togenerate a RF signal for radiation to the environment with reducedsignal distortions and with reduced power consumption.

Predistortion circuit 2310 may further support different/multipleconfigurations as described above. Accordingly, wireless transceiver2300 comprises a feedback path from antenna 2390 to adaption circuitry2315. The feedback path receives a portion of the distorted PA outputsignal (e.g. by means of a coupler). The feedback signal isdown-converted to the baseband domain and subsequently digitized by ADC2325. Feedback receiver 2335 further processes the feedback signal. Theadaption circuitry 2315 updates a predistortion parameter (e.g. apredistortion function) of the presently used predistorter configurationby comparing a sequence of the processed feedback signal with a timealigned sequence of baseband signal 2301 as described above. Thefeedback signal may alternatively or additionally be provided by theregular receive path 2395 of wireless transceiver 2300.

In order to summarize the above aspects on downsampling MDPD, FIG. 24further illustrates a flowchart of a method 2400 for predistorting abaseband signal. Method 2400 comprises determining 2402 samples of thebaseband signal at a first sample rate. Further, method 2400 comprisescalculating 2404 samples of a predistorted baseband signal at a secondsample rate based on predistortion parameters and the samples of thebaseband signal. The second sample rate is lower than the first samplerate. Method 2400 additionally comprises calculating 2406 samples of thepredistorted baseband signal at the first sample rate based on thesamples of the predistorted baseband signal at the second sample rate.

Since the input sample rate of the MDPD is high, amplitude values of thebaseband signal may be tracked with high accuracy. Further, since theoutput sample rate of the MDPD is low, the MDPD calculates only therequired output samples, which results in low current consumption.Accordingly, the MDPD achieves both high accuracy and low currentconsumption at the same time.

More details and aspects of the method are mentioned in connection withthe proposed technique or one or more examples described above (e.g.FIGS. 16 to 23). The method may comprise one or more additional optionalfeatures corresponding to one or more aspects of the proposed techniqueor one or more examples described above.

An example of an implementation using predistortion according to one ormore aspects of the proposed technique or one or more examples describedabove is illustrated in FIG. 25. FIG. 25 schematically illustrates anexample of a mobile device 2500 (e.g. mobile phone, smartphone,tablet-computer, or laptop).

Mobile device 2500 comprises a wireless transmitter or transmittercircuit 2510 according to one or more aspects of the proposed techniqueor one or more examples described above. At least one antenna 2260 ofthe mobile device 2500 is coupled to the wireless transmitter ortransmitter circuit 2510.

The wireless transmitter or transmitter circuit 2510 may comprise apredistortion circuit 2520, a control circuit 2530 for predistortioncircuit 2520 and/or an apparatus for predistorting a baseband signal2540 according to one or more aspects of the proposed technique or oneor more examples described above. Outputs of the predistortion circuit2520 and/or the apparatus for predistorting a baseband signal 2540 maybe coupled to a PA 2550 for amplifying the predistorted signals.

To this end, a mobile device enabling generation of high bandwidthtransmit signals with reduced current consumption may be provided.

The above wireless communication circuits using predistortion accordingto the proposed technique or one or more of the examples described abovemay be configured to operate according to one of the 3^(rd) GenerationPartnership Project (3GPP)-standardized mobile communication networks orsystems. The mobile or wireless communication system may correspond to,for example, a 5G New Radio (5G NR), a Long-Term Evolution (LTE), anLTE-Advanced (LTE-A), High Speed Packet Access (HSPA), a UniversalMobile Telecommunication System (UMTS) or a UMTS Terrestrial RadioAccess Network (UTRAN), an evolved-UTRAN (e-UTRAN), a Global System forMobile communication (GSM) or Enhanced Data rates for GSM Evolution(EDGE) network, a GSM/EDGE Radio Access Network (GERAN). Alternatively,the wireless communication circuits may be configured to operateaccording to mobile communication networks with different standards, forexample, a Worldwide Inter-operability for Microwave Access (WIMAX)network IEEE 802.16 or Wireless Local Area Network (WLAN) IEEE 802.11,generally an Orthogonal Frequency Division Multiple Access (OFDMA)network, a Time Division Multiple Access (TDMA) network, a Code DivisionMultiple Access (CDMA) network, a Wideband-CDMA (WCDMA) network, aFrequency Division Multiple Access (FDMA) network, a Spatial DivisionMultiple Access (SDMA) network, etc.

The examples as described herein may be summarized as follows:

Example 1 is a predistortion circuit for a wireless transmitter,comprising: a signal input configured to receive a baseband signal; anda predistorter configured to generate a predistorted baseband signalusing the baseband signal and a select of one of a first predistorterconfiguration and a second predistorter configuration.

In example 2, the first predistorter configuration in the predistortioncircuit of example 1 performs a first predistortion function and whereinthe second predistorter configuration performs a second predistortionfunction.

In example 3, the predistortion circuit of example 1 or 2 furthercomprises: a first number of computation nodes active in the firstpredistorter configuration; and a second number of computation nodesactive in the second predistorter configuration.

In example 4, the second number in the predistortion circuit of example3 is higher than the first number.

In example 5, the first number of computation nodes in the predistortioncircuit of example 3 are configured to operate at a first rate, whereinthe second number of computation nodes are configured to operate at asecond rate, and wherein the second rate is higher than the first rate.

In example 6, the predistortion circuit of example 5 further comprises asample rate converter configured to match an output of the first numberof computation nodes to the second rate to generate a rate matchedoutput of the first number of computation nodes.

In example 7, the predistortion circuit of example 6 further comprises acombination circuit configured to combine an output of the second numberof computation nodes and the rate matched output of the first number ofcomputation nodes.

In example 8, the predistortion circuit of any of the preceding examplesfurther comprises a configuration handling circuit configured to selectthe first predistorter configuration or the second predistorterconfiguration depending on an operating characteristic of the wirelesstransmitter.

In example 9, the operating characteristic in the predistortion circuitof example 8 comprises at least one of an Average Power Tracking Mode,an Envelope Tracking Mode, an output power range, a peak-to-averagepower ratio of the baseband signal, a modulation scheme used to generatethe baseband signal, and a matching condition of an antenna.

In example 10, the operating characteristic in the predistortion circuitof example 8 or 9 comprises at least one of a transmit bandwidth, atransmit band, a transmit frequency range within the transmit band, anumber of transmit clusters in a frequency domain, a frequencyseparation between the transmit clusters, and a bandwidth of eachtransmit cluster.

In example 11, the operating characteristic in the predistortion circuitof any of examples 8 to 10 comprises an acceptable spectral mask.

In example 12, the configuration handling circuit in the predistortioncircuit of example 8 further comprises an input interface configured toreceive a feedback signal depending on an output of a power amplifier ofthe transmitter, wherein the configuration handling circuit is furtherconfigured to select the first predistorter configuration or the secondpredistorter configuration depending on the feedback signal.

Example 13 is a predistortion circuit for a wireless transmitter,comprising: a signal input configured to receive a baseband signal; anda predistorter configured to generate a predistorted baseband signalusing a first sub-circuit to calculate a first portion of a predistortedbaseband signal at a first rate and a second sub-circuit to calculate afirst portion of a predistorted baseband signal at a second rate.

In example 14, the predistortion circuit of example 13 further comprisesrate matching circuitry configured to adjust the first rate and thesecond rate to a sampling rate of the predistorted baseband signal.

In example 15, the predistortion circuit of example 13 or 14 furthercomprises a combination circuit configured to combine the first portionof the predistorted baseband signal and the second portion of thepredistorted baseband signal to generate the predistorted basebandsignal.

Example 16 is a method for generating a predistorted baseband signal fora wireless transmitter, comprising: selecting a first predistorterconfiguration or a second predistorter configuration; and generating thepredistorted baseband signal using a baseband signal and the selectpredistorter configuration.

In example 17, the method of example 16 further comprises: performing afirst predistortion function in the first predistorter configuration;and performing a second predistortion function in the secondpredistorter configuration.

In example 18, the method of example 16 or 17 further comprises: using afirst number of computations in the first predistorter configuration;and using a second number of computations in the second predistorterconfiguration.

In example 19, the second number is higher than the first number in themethod of example 18.

In example 20, the method of example 18 further comprises: performingthe first number of computations at a first rate; and performing thesecond number of computations at a second rate, the first rate beinghigher than the second rate.

In example 21, the method of example 18 further comprises matching anoutput of the first number of computations to the second rate togenerate a rate matched output of the first number of computation nodes.

In example 22, the method of example 21 further comprises combining anoutput of the second number of computations with the rate matched outputof the first number of computations.

In example 23, the method of any of examples 16 to 22 further comprisesselecting the first predistorter configuration or the secondpredistorter configuration depending on an operating characteristic ofthe wireless transmitter.

In example 24, the operating characteristic in the method of example 23comprises at least one of an Average Power Tracking Mode, an EnvelopeTracking Mode, an output power range, a peak-to-average power ratio ofthe baseband signal, a modulation scheme used to generate the basebandsignal, and a matching condition of an antenna.

In example 25, the operating characteristic in the method of example 23or 24 comprises at least one of a transmit bandwidth, a transmit band, atransmit frequency range within the transmit band, a number of transmitclusters in a frequency domain, a frequency separation between thetransmit clusters, and a bandwidth of each transmit cluster.

In example 26, the operating characteristic in the method of any ofexamples 23 to 25 comprises an acceptable spectral mask.

In example 27, the method of any of examples 16 to 26 further comprises:receiving a feedback signal depending on an output of a power amplifierof the transmitter; and selecting the first predistorter configurationor the second predistorter configuration depending on the feedbacksignal.

Example 28 is a wireless transmitter comprising a predistortion circuitaccording to any one of examples 1 to 15.

In example 29, the wireless transmitter of example 28 further comprisesa power amplifier coupled to an output of the predistortion circuit.

Example 30 is a mobile device comprising a wireless transmitteraccording to one of examples 28 or 29.

In example 31, the mobile device of example 30 further comprises atleast one antenna coupled to the wireless transmitter.

Example 32 is a non-transitory computer readable medium having storedthereon a program having a program code for performing the method of anyof examples 16 to 27, when the program is executed on a computer orprocessor.

Example 33 is a computer program having a program code configured toperform the method of any of examples 16 to 27, when the computerprogram is executed on a computer or processor.

Example 34 is a control circuit for a predistortion circuit, the controlcircuit comprising: a feedback signal input configured to receive afeedback signal related to an output of a power amplifier; bandwidthlimitation circuitry configured to limit a bandwidth of the feedbacksignal to derive a bandlimited feedback signal; and a parameter handlingcircuit configured to update predistortion parameters used within thepredistortion circuit based on the bandlimited feedback signal.

In example 35, the control circuit of example 34 further comprises anoutput interface configured to output the updated predistortionparameters.

In example 36, the bandwidth limitation circuitry in the control circuitof example 34 or 35 comprises: an adjustable mixer configured to downmixthe feedback signal using a feedback oscillator signal to generate adownmixed feedback signal, the feedback oscillator signal having afrequency of a local oscillator signal plus an observation frequencyoffset; and a band pass filter circuit configured to band filter thedownmixed feedback signal to generate the bandlimited feedback signal.

In example 37, the control circuit of any of examples 34 to 36 furthercomprises a reference signal generation circuit comprising: a referencesignal input configured to receive a reference signal related to anoutput of the predistortion circuit or related to an input of thepredistortion circuit; and further bandwidth limitation circuitryconfigured to limit a bandwidth of the reference signal to derive abandlimited reference signal, wherein the configuration handling circuitis configured to update the predistortion parameters based on thebandlimited observation signal and the bandlimited reference signal.

In example 38, the further bandwidth limitation circuitry in the controlcircuit of example 37 comprises: a frequency shifter configured to shiftthe reference signal by a frequency offset; and a further band passfilter circuit configured to band filter the shifted reference signal togenerate the bandlimited feedback signal.

In example 39, the frequency offset in the control circuit of example 38corresponds to an observation frequency offset of a feedback signal.

Example 40 is a transmitter circuit, comprising: a predistortion circuitconfigured to receive a baseband signal and to generate a predistortedbaseband signal based on predistortion parameters; an upmixer configuredto generate a radio frequency signal using the predistorted basebandsignal and a local oscillator signal; a power amplifier coupled to anoutput of the upmixer and configured to generate an amplified radiofrequency signal; and a control circuit for the predistortion circuit,the control circuit comprising: a feedback signal input configured toreceive a feedback signal related to an output of a power amplifier;bandwidth limitation circuitry configured to limit a bandwidth of thefeedback signal to derive a bandlimited feedback signal; a referencesignal input configured to a receive reference signal related to anoutput of the predistortion circuit or related to an input of thepredistortion circuit; further bandwidth limitation circuitry configuredto limit a bandwidth of the reference signal to derive a bandlimitedreference signal; and a parameter handling circuit configured to updatethe predistortion parameters used within the predistortion circuit basedon the bandlimited feedback signal and on the bandlimited referencesignal.

In example 41, the bandwidth of the bandlimited feedback signal and ofthe bandlimited reference signal are equal in the transmitter circuit ofexample 40.

In example 42, the control circuit in the transmitter circuit of example40 or 41 further comprises a phase control input coupled to an output ofthe upmixer to receive a phase control signal, wherein the controlcircuit is further configured to adjust a phase of the feedback signalbased on the phase control signal.

In example 43, the bandwidth limitation circuitry in the transmittercircuit of any of examples 40 to 42 further comprises an adjustablemixer configured to downmix the feedback signal using a feedbackoscillator signal, the feedback oscillator signal having an observationfrequency offset from the frequency of the local oscillator signal; andthe further bandwidth limitation circuitry comprises a frequency shifterconfigured to shift the reference signal by the inverse of theobservation frequency offset.

Example 44 is a mobile device comprising a transmitter circuit accordingto one of examples 40 to 43.

In example 40, the mobile device of example 40 further comprises atleast one antenna coupled to the transmitter circuit.

Example 46 is a method to determine parameters for a predistortioncircuit, the method comprising: receiving a feedback signal related toan output of a power amplifier; limiting a bandwidth of the feedbacksignal to derive a bandlimited feedback signal; and updatingpredistortion parameters used within the predistortion circuit based onthe bandlimited feedback signal.

In example 47, the method of example 46 further comprises: downmixingthe feedback signal to generate a downmixed feedback signal using afeedback oscillator signal to generate a downmixed feedback signal, thefeedback oscillator signal having a frequency of a local oscillatorsignal plus an observation frequency offset; and band pass filtering thedownmixed feedback signal.

In example 48, the method of example 46 or 47 further comprises:receiving a reference signal related to related to an output of thepredistortion circuit or related to an input of the predistortioncircuit; limiting a bandwidth of the reference signal to derive abandlimited reference signal; and updating the predistortion parametersbased on the bandlimited observation signal and the bandlimitedreference signal.

In example 49, the method of any of examples 46 to 48 further comprisesshifting the reference signal by a frequency offset.

In example 50, the frequency offset in the method of example 49corresponds to an observation frequency offset of a feedback signal.

Example 51 is a non-transitory computer readable medium having storedthereon a program having a program code for performing the method of anyof examples 46 to 50, when the program is executed on a computer orprocessor.

Example 52 is a computer program having a program code configured toperform the method of any of examples 46 to 50, when the computerprogram is executed on a computer or processor.

Example 53 is an apparatus for predistorting a baseband signal,comprising: a predistortion circuit configured to determine samples ofthe baseband signal at a first sample rate and calculate samples of apredistorted baseband signal at a second sample rate based onpredistortion parameters and the samples of the baseband signal, whereinthe second sample rate is lower than the first sample rate; and anupsampling filter configured to calculate samples of the predistortedbaseband signal at the first sample rate based on the samples of thepredistorted baseband signal at the second sample rate.

In example 54, the first sample rate in the apparatus of example 53 isequal to or higher than a sample rate of the baseband signal.

In example 55, the predistortion circuit in the apparatus of example 53or example 54 is further configured to: receive information about aspectral allocation of the baseband signal's data in a frequencyspectrum; calculate the samples of the predistorted baseband signal atthe second sample rate, if the spectral allocation satisfies a firstdecision criterion; and calculate the samples of the predistortedbaseband signal at the first sample rate, if the spectral allocationsatisfies a second decision criterion.

In example 56, the upsampling filter in the apparatus of example 55 isdeactivated, if the spectral allocation satisfies the second decisioncriterion.

Example 57 is a wireless transmitter comprising an apparatus forpredistorting a baseband signal according to any one of examples 53 to56.

In example 58, the wireless transmitter of example 57 further comprisesa power amplifier coupled to an output of the apparatus forpredistorting a baseband signal.

Example 59 is a mobile device comprising a wireless transmitteraccording to examples 57 or example 58.

In example 60, the mobile device of example 59 further comprises atleast one antenna coupled to the wireless transmitter.

Example 61 is a method for predistorting a baseband signal, comprising:determining samples of the baseband signal at a first sample rate;calculating samples of a predistorted baseband signal at a second samplerate based on predistortion parameters and the samples of the basebandsignal, wherein the second sample rate is lower than the first samplerate; and calculating samples of the predistorted baseband signal at thefirst sample rate based on the samples of the predistorted basebandsignal at the second sample rate.

In example 62, the first sample rate in the method of example 61 isequal to or higher than a sample rate of the baseband signal.

In example 63, the method of example 61 or example 62 further comprises:receiving information about a spectral allocation of the basebandsignal's data in a frequency spectrum, wherein the samples of thepredistorted baseband signal are calculated at the second sample rate,if the spectral allocation satisfies a first decision criterion; andcalculating the samples of the predistorted baseband signal at the firstsample rate, if the spectral allocation satisfies a second decisioncriterion.

In example 64, an upsampling filter is used in the method of example 63for calculating the samples of the predistorted baseband signal at thefirst sample rate based on the samples of the predistorted basebandsignal at the second sample rate, wherein the method further comprisesdeactivating the upsampling filter, if the spectral allocation satisfiesthe second decision criterion.

Example 65 is a non-transitory computer readable medium having storedthereon a program having a program code for performing the method of anyof examples 61 to 64, when the program is executed on a computer orprocessor.

Example 66 is a computer program having a program code configured toperform the method of any of examples 61 to 64, when the program isexecuted on a computer or processor.

The aspects and features mentioned and described together with one ormore of the previously detailed examples and figures, may as well becombined with one or more of the other examples in order to replace alike feature of the other example or in order to additionally introducethe feature to the other example.

Examples may further be or relate to a computer program having a programcode for performing one or more of the above methods, when the computerprogram is executed on a computer or processor. Steps, operations orprocesses of various above-described methods may be performed byprogrammed computers or processors. Examples may also cover programstorage devices such as digital data storage media, which are machine,processor or computer readable and encode machine-executable,processor-executable or computer-executable programs of instructions.The instructions perform or cause performing some or all of the acts ofthe above-described methods. The program storage devices may comprise orbe, for instance, digital memories, magnetic storage media such asmagnetic disks and magnetic tapes, hard drives, or optically readabledigital data storage media. Further examples may also cover computers,processors or control units programmed to perform the acts of theabove-described methods or (field) programmable logic arrays ((F)PLAs)or (field) programmable gate arrays ((F)PGAs), programmed to perform theacts of the above-described methods.

The description and drawings merely illustrate the principles of thedisclosure. Furthermore, all examples recited herein are principallyintended expressly to be only for pedagogical purposes to aid the readerin understanding the principles of the disclosure and the conceptscontributed by the inventor(s) to furthering the art. All statementsherein reciting principles, aspects, and examples of the disclosure, aswell as specific examples thereof, are intended to encompass equivalentsthereof.

A functional block denoted as “means for . . . ” performing a certainfunction may refer to a circuit that is configured to perform a certainfunction. Hence, a “means for s.th.” may be implemented as a “meansconfigured to or suited for s.th.”, such as a device or a circuitconfigured to or suited for the respective task.

Functions of various elements shown in the figures, including anyfunctional blocks labeled as “means”, “means for providing a signal”,“means for generating a signal.”, etc., may be implemented in the formof dedicated hardware, such as “a signal provider”, “a signal processingunit”, “a processor”, “a controller”, etc. as well as hardware capableof executing software in association with appropriate software. Whenprovided by a processor, the functions may be provided by a singlededicated processor, by a single shared processor, or by a plurality ofindividual processors, some of which or all of which may be shared.However, the term “processor” or “controller” is by far not limited tohardware exclusively capable of executing software, but may includedigital signal processor (DSP) hardware, network processor, applicationspecific integrated circuit (ASIC), field programmable gate array(FPGA), read only memory (ROM) for storing software, random accessmemory (RAM), and non-volatile storage. Other hardware, conventionaland/or custom, may also be included.

A block diagram may, for instance, illustrate a high-level circuitdiagram implementing the principles of the disclosure. Similarly, a flowchart, a flow diagram, a state transition diagram, a pseudo code, andthe like may represent various processes, operations or steps, whichmay, for instance, be substantially represented in computer readablemedium and so executed by a computer or processor, whether or not suchcomputer or processor is explicitly shown. Methods disclosed in thespecification or in the claims may be implemented by a device havingmeans for performing each of the respective acts of these methods.

It is to be understood that the disclosure of multiple acts, processes,operations, steps or functions disclosed in the specification or claimsmay not be construed as to be within the specific order, unlessexplicitly or implicitly stated otherwise, for instance for technicalreasons. Therefore, the disclosure of multiple acts or functions willnot limit these to a particular order unless such acts or functions arenot interchangeable for technical reasons. Furthermore, in some examplesa single act, function, process, operation or step may include or may bebroken into multiple sub-acts, -functions, -processes, -operations or-steps, respectively. Such sub acts may be included and part of thedisclosure of this single act unless explicitly excluded.

Furthermore, the following claims are hereby incorporated into thedetailed description, where each claim may stand on its own as aseparate example. While each claim may stand on its own as a separateexample, it is to be noted that—although a dependent claim may refer inthe claims to a specific combination with one or more other claims—otherexamples may also include a combination of the dependent claim with thesubject matter of each other dependent or independent claim. Suchcombinations are explicitly proposed herein unless it is stated that aspecific combination is not intended. Furthermore, it is intended toinclude also features of a claim to any other independent claim even ifthis claim is not directly made dependent to the independent claim.

1. A predistortion circuit comprising: a first sub-circuit configured tocalculate a first part of a predistorted baseband signal based on areceived baseband signal and generate a corresponding first predistortedbaseband signal having a first rate, and a second sub-circuit configuredto calculate a second part of the predistorted baseband based on thereceived baseband signal and generate a corresponding secondpredistorted baseband signal having a second rate, wherein thepredistortion circuit is configured to produce the predistorted basebandsignal based one or more of the first predistorted baseband signal andthe second predistorted baseband signal.
 2. The predistortion circuit ofclaim 1, wherein the first sub-circuit and the second sub-circuit arecoupled in series, wherein an output of the first sub-circuit is coupledto an input of the second sub-circuit.
 3. The predistortion circuit ofclaim 2, wherein the second predistorted baseband signal is thepredistorted baseband signal.
 4. The predistortion circuit of claim 2,wherein the second sub-circuit comprises a sample rate converter havingan input coupled to the output of the first sub-circuit, wherein thesample rate converter is configured to convert the first rate to thesecond rate.
 5. The predistortion circuit of claim 2, further comprisingone or more additional sub-circuits coupled in series between the outputof the first sub-circuit and the input of the second sub-circuit,wherein each sub-circuit of the one or more additional sub-circuits isconfigured to calculate a different respective part of the predistortedbaseband signal and generate a corresponding respective predistortedbaseband signal having a different corresponding rate.
 6. Thepredistortion circuit of claim 5, wherein each respective sub-circuit ofthe one or more additional sub-circuits comprises a respective rateconverter configured to: receive a corresponding signal output from apreceding sub-circuit coupled to the respective sub-circuit in theseries; and convert the respective rate of the corresponding signal tothe respective rate corresponding to the respective sub-circuit.
 7. Thepredistortion circuit of claim 1, wherein the first rate and the secondrate respectively correspond to different signal bandwidths.
 8. Thepredistortion circuit of claim 1, further comprising: rate matchingcircuitry configured to equalize the first rate of the firstpredistorted baseband signal and the second rate of the secondpredistorted baseband signal to a rate of the predistorted basebandsignal.
 9. The predistortion circuit of claim 8, wherein the firstsub-circuit and the second sub-circuit are coupled in parallel, whereinthe predistortion circuit further comprises: a combining circuitconfigured to combine the rate-equalized first predistorted basebandsignal and the rate-equalized second predistorted baseband signal intothe predistorted baseband signal.
 10. A method for generating apredistorted baseband signal, the method comprising: calculating a firstpart of a predistorted baseband signal based on a received basebandsignal and generating a corresponding first predistorted baseband signalhaving a first rate; calculating a second part of the predistortedbaseband signal based on the received baseband signal and generating acorresponding second predistorted baseband signal having a second rate;and producing the predistorted baseband signal based on one or more ofthe first predistorted baseband signal and the second predistortedbaseband signal.
 11. The method of claim 10, wherein producing thepredistorted baseband signal comprises: generating the secondpredistorted baseband signal based on the first predistorted basebandsignal; and providing the second predistorted baseband signal as thepredistorted baseband signal.
 12. The method of claim 10 furthercomprising: converting the first rate to the second rate subsequent tocalculating the first part of the predistorted baseband signal and priorto calculating the second part of the predistorted baseband signal. 13.The method of claim 10, wherein producing the predistorted basebandsignal comprises equalizing the first rate of the first predistortedbaseband signal and the second rate of the second predistorted basebandsignal to a rate of the predistorted baseband signal.
 14. The method ofclaim 13, wherein producing the predistorted baseband signal furthercomprises combining the rate-equalized first predistorted basebandsignal and the rate-equalized second predistorted baseband signal.
 15. Awireless device comprising: at least one antenna; and a transmittercircuit coupled to the at least one antenna and configured to calculatea first part of a predistorted baseband signal based on a receivedbaseband signal and generate a corresponding first predistorted basebandsignal having a first rate, calculate a second part of the predistortedbaseband signal based on the received baseband signal and generate acorresponding second predistorted baseband signal having a second rate,and produce the predistorted baseband signal based on one or more of thefirst predistorted baseband signal and the second predistorted basebandsignal.
 16. The wireless device of claim 15, wherein the transmittercircuit is further configured to: generate the second predistortedbaseband signal based on the first predistorted baseband signal; andprovide the second predistorted baseband signal as the predistortedbaseband signal.
 17. The wireless device of claim 15, wherein thetransmitter circuit is further configured to convert the first rate tothe second rate subsequent to calculating the first part of thepredistorted baseband signal and prior to calculating the second part ofthe predistorted baseband signal.
 18. The wireless device of claim 15,wherein the transmitter circuit is further configured to equalize thefirst rate of the first predistorted baseband signal and the second rateof the second predistorted baseband signal to a rate of the predistortedbaseband signal prior to producing the predistorted baseband signal. 19.The wireless device of claim 18, wherein the transmitter circuit isfurther configured to combine the rate-equalized first predistortedbaseband signal and the rate equalized second predistorted basebandsignal to produce the predistorted baseband signal.
 20. The wirelessdevice of claim 15, further comprising: a power amplifier configured toreceive a radio frequency version of the predistorted baseband signaland amplify the radio frequency version of the predistorted basebandsignal for transmission via the at least one antenna.